LT8304/LT8304-1 100VIN Micropower No-Opto Isolated Flyback Converter with 150V/2A Switch FEATURES DESCRIPTION 3V to 100V Input Voltage Range nn 2A, 150V Internal DMOS Power Switch nn Low Quiescent Current: nn 116A in Sleep Mode nn 390A in Active Mode nn Quasi-Resonant Boundary Mode Operation at Heavy Load nn Low Ripple Burst Mode(R) Operation at Light Load nn Minimum Load < 0.5% (Typ) of Full Output nn No Transformer Third Winding or Opto-Isolator Required for Output Voltage Regulation nn Accurate EN/UVLO Threshold and Hysteresis nn Internal Compensation and Soft-Start nn Temperature Compensation for Output Diode nn Output Short-Circuit Protection nn Thermally Enhanced 8-Lead SO Package The LT(R)8304/LT8304-1 are monolithic micropower isolated flyback converters. By sampling the isolated output voltage directly from the primary-side flyback waveform, the parts require no third winding or opto-isolator for regulation. The output voltage is programmed with two external resistors and a third optional temperature compensation resistor. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 2A, 150V DMOS power switch is integrated along with all the high voltage circuitry and control logic into a thermally enhanced 8-lead SO package. nn APPLICATIONS Isolated Automotive, Industrial, Medical, Telecom Power Supplies nn Isolated Auxiliary/Housekeeping Power Supplies nn The LT8304/LT8304-1 operate from an input voltage range of 3V to 100V and deliver up to 24W of isolated output power. The high level of integration and the use of boundary and low ripple Burst Mode operation result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. The LT8304-1 is specially optimized for high step-up output applications. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522. TYPICAL APPLICATION 4V to 80VIN/5VOUT Isolated Flyback Converter 6:1 220pF 10F 40H 100 VIN SW EN/UVLO LT8304 GND 1F INTVCC * * VOUT+ 5V 1.1H VOUT- 309k RFB 20mA TO 2.4A (VIN = 24V) 20mA TO 3.6A (VIN = 48V) 20mA TO 4.2A (VIN = 72V) RREF 100k TC 90 100F x3 EFFICIENCY (%) VIN 4V TO 80V Efficiency vs Load Current 100 80 70 60 10k 8304 TA01a VIN = 24V VIN = 48V VIN = 72V 50 40 0 0.6 1.2 1.8 2.4 3.0 LOAD CURRENT (A) 3.6 4.2 8304 TA01b 8304fa For more information www.linear.com/LT8304 1 LT8304/LT8304-1 ABSOLUTE MAXIMUM RATINGS (Note 1) SW (Note 2).............................................................150V VIN...........................................................................100V EN/UVLO.....................................................................VIN RFB.........................................................VIN - 0.5V to VIN Current Into RFB.....................................................200A INTVCC, RREF, TC..........................................................4V Operating Junction Temperature TJ Range (Notes 3, 4) LT8304E/LT8304E-1........................... -40C to 125C LT8304I/LT8304I-1............................. -40C to 125C LT8304H/LT8304H-1.......................... -40C to 150C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec).................... 300C ORDER INFORMATION PIN CONFIGURATION TOP VIEW EN/UVLO 1 INTVCC 2 VIN 3 GND 4 8 9 GND TC 7 RREF 6 RFB 5 SW S8E PACKAGE 8-LEAD PLASTIC SO JA = 33C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB http://www.linear.com/product/LT8304#orderinfo LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8304ES8E#PBF LT8304ES8E#TRPBF 8304 8-Lead Plastic SO -40C to 125C LT8304IS8E#PBF LT8304IS8E#TRPBF 8304 8-Lead Plastic SO -40C to 125C LT8304HS8E#PBF LT8304HS8E#TRPBF 8304 8-Lead Plastic SO -40C to 150C LT8304ES8E-1#PBF LT8304ES8E-1#TRPBF 83041 8-Lead Plastic SO -40C to 125C LT8304IS8E-1#PBF LT8304IS8I-1#TRPBF 83041 8-Lead Plastic SO -40C to 125C LT8304HS8E-1#PBF LT8304HS8E-1#TRPBF 83041 8-Lead Plastic SO -40C to 150C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 2 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, VEN/UVLO = VIN, CINTVCC = 1F to GND, unless otherwise noted. SYMBOL PARAMETER VIN VIN Voltage Range IQ VIN Quiescent Current CONDITIONS MIN l TYP 3 VEN/UVLO = 0.2V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 1.8 63 116 390 EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.2 0.5 EN/UVLO Enable Threshold Falling l 1.178 1.214 EN/UVLO Enable Hysteresis MAX UNIT 100 V 3 A A A A V 1.250 14 V mV IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V VEN/UVLO = 1.3V -0.1 2.3 -0.1 0 2.5 0 0.1 2.7 0.1 A A A VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.8 3 3.1 V IINTVCC INTVCC Current Limit VINTVCC = 2.8V INTVCC UVLO Threshold Falling 16 2.38 INTVCC UVLO Hysteresis (RFB - VIN) Voltage VTC TC Pin Voltage ITC TC Pin Current 2.56 105 IRFB = 75A to 125A -60 RREF Regulation Voltage RREF Regulation Voltage Line Regulation 2.47 mA l 0.98 3V VIN 100V 60 1.02 V 0.02 0.1 % 15 10 -200 18 13 A A A 315 350 385 kHz 8 11 14 kHz fMAX Maximum Switching Frequency Minimum Switching Frequency tON(MIN) Minimum Switch-On Time ISW(MAX) Maximum Switch Current Limit ISW(MIN) Minimum Switch Current Limit RDS(ON) Switch On-Resistance ISW = 0.8A 0.5 ILKG Switch Leakage Current VSW = 150V 0.1 tSS Soft-Start Timer (LT8304) (LT8304-1) 160 950 ns ns 2.0 2.4 2.8 0.43 0.48 0.53 11 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 150V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 150V as shown in Figure 5. Note 3: The LT8304E/LT8304E-1 are guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. V 12 7 fMIN l mV 1.00 1.00 VTC = 1.2V (LT8304) VTC = 1.2V (LT8304-1) VTC = 0.8V V mV A A 0.5 A ms The LT8304I/LT8304I-1 are guaranteed over the full -40C to 125C operating junction temperature range. LT8304H/LT304H-1 are guaranteed over the full -40C to 150C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125C. Note 4: The LT8304/LT8304-1 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 8304fa For more information www.linear.com/LT8304 3 LT8304/LT8304-1 TYPICAL PERFORMANCE CHARACTERISTICS 5.20 5.3 FRONT PAGE APPLICATION 5.05 5.00 4.95 4.90 VIN = 24V VIN = 48V VIN = 72V 4.80 0 0.6 1.2 1.8 2.4 3.0 LOAD CURRENT (A) 3.6 500 FRONT PAGE APPLICATION VIN = 48V IOUT = 1A 5.2 5.10 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 5.15 4.85 5.1 RTC = 100k 5.0 RTC = OPEN 4.9 4.7 -50 -25 0 VSW 50V/DIV VOUT 50mV/DIV VOUT 50mV/DIV 8304 G04 0.6 1.2 1.8 2.4 3 LOAD CURRENT (A) 3.6 Burst Mode Operation Waveforms 8304 G05 8304 G06 20s/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 20mA VIN Quiescent Current, Active Mode 150 TJ = 150C TJ = 25C TJ = -50C 4.2 8304 G03 VIN Quiescent Current, Sleep Mode 450 TJ = 150C 140 TJ = 150C 430 4 2 IQ (A) 130 IQ (A) IQ (A) 0 VOUT 50mV/DIV TJ = 25C 120 110 4 VIN = 24V VIN = 48V VIN = 72V VSW 50V/DIV 2s/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 0.5A VIN Shutdown Current 6 0 0 25 50 75 100 125 150 TEMPERATURE () Discontinuous Mode Waveforms VSW 50V/DIV 8 200 8304 G02 Boundary Mode Waveforms 10 300 100 4.8 4.2 FRONT PAGE APPLICATION 400 8304 G01 2s/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 3A Switching Frequency vs Load Current Output Temperature Variation FREQUENCY (kHz) Output Load and Line Regulation TA = 25C, unless otherwise noted. 20 40 60 VIN (V) 80 100 8304 G07 90 TJ = 25C TJ = -50C 390 TJ = -50C 370 100 0 410 0 20 40 60 VIN (V) 80 100 8304 G08 350 0 20 40 60 VIN (V) 80 100 8304 G09 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 TYPICAL PERFORMANCE CHARACTERISTICS EN/UVLO Enable Threshold TA = 25C, unless otherwise noted. EN/UVLO Hysteresis Current 1.240 3.10 5 1.235 FALLING 1.215 1.210 3 VINTVCC (V) 1.220 2 1 1.200 -50 -25 0 0 8304 G10 2.8 3.05 INTV INTVCC UVLO Threshold Threshold CC UVLO 40 (RFB -- VVIN Voltage IN)) Voltage 30 RISING 2.5 FALLING 2.4 2.85 2.3 VOLTAGE (mV) IINTVCC = 10mA 2.6 2.90 IRFB = 125A 10 IRFB = 100A 0 -10 -20 20 40 60 VIN (V) 80 2.2 -50 -25 100 1.010 1.008 1.008 1.006 1.006 1.004 1.004 1.002 1.002 1.000 0.998 0.994 0.994 0.992 0.992 8304 G16 25 50 75 100 125 150 TEMPERATURE (C) TC Pin Voltage 1.5 1.4 1.3 1.2 0.998 0.996 25 50 75 100 125 150 TEMPERATURE (C) 0 8304 G15 RREF REF Line Regulation 1.000 0.996 0 -40 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 8304 G14 R RREF Regulation Voltage Voltage REF Regulation 0.990 -50 -25 0 8304 G13 VRREF (V) VRREF (V) 1.010 0 IRFB = 75A -30 VTC (V) 2.80 25 50 75 100 125 150 TEMPERATURE (C) 20 IINTVCC = 0mA 2.95 0 8304 G12 2.7 VINTVCC (V) VINTVCC (V) 2.80 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 8304 G11 INTVCC Voltage vs VIN 3.00 IINTVCC = 10mA 2.85 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 2.95 2.90 1.205 3.10 IINTVCC = 0mA 3.00 1.225 IHYST (A) VEN/UVLO (V) 3.05 4 RISING 1.230 INTVCC Voltage vs Temperature 0.990 1.1 1.0 0.9 0.8 0 20 40 60 VIN (V) 80 100 8304 G17 0.7 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 8304 G18 8304fa For more information www.linear.com/LT8304 5 LT8304/LT8304-1 TYPICAL PERFORMANCE CHARACTERISTICS RDS(ON) DS(ON) Switch Current Limit 2.5 0.8 2.0 0.6 400 1.5 0.4 1.0 0.2 0.5 0 MAXIMUM CURRENT LIMIT FREQUENCY (kHz) 1.0 0 -50 -25 MINIMUM CURRENT LIMIT 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 8304 G19 Minimum Switching Frequency Minimum Switch-Off Time 400 TIME (ns) FREQUENCY (kHz) TIME (ns) 25 50 75 100 125 150 TEMPERATURE (C) 8304 G22 25 50 75 100 125 150 TEMPERATURE (C) 500 200 100 0 0 8304 G21 300 4 0 -50 -25 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) Minimum Switch-On Time 16 8 200 100 400 12 300 8304 G20 20 6 Maximum Switching Frequency 500 3.0 ISW (A) RESISTANCE () 1.2 TA = 25C, unless otherwise noted. 0 -50 -25 300 200 100 0 25 50 75 100 125 150 TEMPERATURE (C) 8304 G23 0 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 8304 G24 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 PIN FUNCTIONS EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8304. Pull the pin below 0.2V to shut down the LT8304. This pin has an accurate 1.214V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from VIN to ground. A 2.5A current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. INTVCC (Pin 2): Internal 3V Linear Regulator Output. The INTVCC pin is supplied from VIN and powers the internal control circuitry and gate driver. Do not overdrive the INTVCC pin with any external supply, such as a third winding supply. Locally bypass this pin to ground with a minimum 1F ceramic capacitor. VIN (Pin 3): Input Supply. The VIN pin supplies current to the internal circuitry and serves as a reference voltage for the feedback circuitry connected to the RFB pin. Locally bypass this pin to ground with a capacitor. GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed pad provides both electrical contact to ground and good thermal contact to the printed circuit board. Solder the exposed pad directly to the ground plane. SW (Pin 5): Drain of the Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. RFB (Pin 6): Input Pin for External Feedback Resistor. Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the RREF resistor, times the internal voltage reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. RREF (Pin 7): Input Pin for External Ground Referred Reference Resistor. The resistor at this pin should be in the range of 10k, but for convenience in selecting a resistor divider ratio, the value may range from 9.09k to 11.0k. TC (Pin 8): Output Voltage Temperature Compensation. The voltage at this pin is proportional to absolute temperature (PTAT) with temperature coefficient equal to 3.35mV/C, i.e., equal to 1V at room temperature 25C. The TC pin voltage can be used to estimate the LT8304 junction temperature. Connect a resistor from this pin to the RREF pin to compensate the output diode temperature coefficient. 8304fa For more information www.linear.com/LT8304 7 LT8304/LT8304-1 BLOCK DIAGRAM T1 N:1 VIN CIN L1A RFB 3 2 6 1:4 M3 M2 REN2 OSCILLATOR - 1.214V + 1V - A1 VOUT- START-UP, REFERENCE, CONTROL BOUNDARY DETECTOR + 1 EN/UVLO COUT SW 25A REN1 L1B VIN LDO CINTVCC * VOUT+ 5 RFB VIN INTVCC * DOUT INTVCC - gm + S A3 R Q M1 DRIVER 2.5A + PTAT VOLTAGE M4 A2 RSENSE - RREF 7 GND 4, EXPOSED PAD PIN 9 TC 8 RTC 8304 BD RREF OPERATION The LT8304 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing 8 extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer's physical size and cost, and dynamic response is often mediocre. The LT8304 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8304 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 OPERATION The LT8304 is a simple to use micropower isolated flyback converter housed in a thermally enhanced 8-lead SO package. The output voltage is programmed with two external resistors. An optional TC resistor provides easy output diode temperature compensation. By integrating the loop compensation and soft-start inside, the part reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sampleand-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Quasi-Resonant Boundary Mode Operation The LT8304 features quasi-resonant boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero and the SW rings to its valley. Boundary conduction mode is a variable frequency, variable peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on at its valley. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit subharmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8304 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 350kHz (TYP). Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8304 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8304 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-andhold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effective quiescent current to improve light load efficiency. In this condition, the LT8304 runs in low ripple Burst Mode operation. The typical 11kHz minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement. High Step-Up VOUT Applications Typically, high step-up output applications have excessive primary inductor current ringing during primary switch turn-on due to the huge reflected capacitance on SW node. Such current ringing can falsely trigger LT8304 current comparator after 160ns typical blanking time and create large signal oscillation, especially at high VIN and light load condition. The LT8304-1, specially optimized for high step-up output applications, is more immune to the current ringing without requiring longer blanking time. For any 1:N step-up transformer turns ratio larger than or equal to 5, the LT8304-1 is recommended. 8304fa For more information www.linear.com/LT8304 9 LT8304/LT8304-1 APPLICATIONS INFORMATION Output Voltage The RFB and RREF resistors as depicted in the Block Diagram are external resistors used to program the output voltage. The LT8304 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the VIN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC * ESR) * NPS VF = Output diode forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current, IRFB, by the RFB resistor and the flyback pulse sense circuit (M2 and M3). This current, IRFB, also flows through the RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sampleand-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC * ESR) term in the VFLBK equation can be assumed to be zero. The internal reference voltage, VREF, 1.00V, feeds to the noninverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes the voltage at the RREF pin to be nearly equal to the internal reference voltage VREF. The resulting relationship between VFLBK and VREF can be expressed as: V FLBK *R = V or R REF REF FB R VFLBK = VREF * FB RREF Combination with the previous VFLBK equation yields an equation for VOUT, in terms of the RFB and RREF resistors, transformer turns ratio, and diode forward voltage: R VOUT = VREF * FB RREF 1 -V * F NPS Output Temperature Compensation The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage, VF, has a significant negative temperature coefficient (-1mV/C to -2mV/C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. The LT8304 junction temperature usually tracks the output diode junction temperature to the first order. To compensate the negative temperature coefficient of the output diode, a resistor, RTC, connected between the TC and RREF pins generates a proportional-to-absolute-temperature (PTAT) current. The PTAT current is zero at 25C, flows into the RREF pin at hot temperature, and flows out of the RREF pin at cold temperature. With the RTC resistor in place, the output voltage equation is revised as follows: VOUT = VREF * RFB 1 * - VF (TO) - ( VTC / T ) * RREF NPS ( T -TO) * RFB 1 * - ( VF / T ) * ( T-TO) R TC NPS TO=Room temperature 25C ( ( VF / T ) = Output diode forward voltage temperature coefficient VTC / T ) = 3.35mV/ C VREF = Internal reference voltage 1.00V 10 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 APPLICATIONS INFORMATION To cancel the output diode temperature coefficient, the following two equations should be satisfied: VOUT = VREF * ( RFB 1 * - VF (TO) RREF NPS R 1 VTC / T) * FB * = - ( VF / T ) R TC NPS Selecting Actual RREF, RFB, RTC Resistor Values The LT8304 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the RFB and RTC resistor values. Therefore, a simple 2-step sequential process is recommended for selecting resistor values. Rearrangement of the expression for VOUT in the previous sections yields the starting value for RFB: RFB = ( RREF *NPS * VOUT + VF (TO) VREF ) VOUT = Output voltage VF (TO) = Output diode forward voltage at 25C = ~0.3V NPS = Transformer effective primary-to-secondary turns ratio The equation shows that the RFB resistor value is independent of the RTC resistor value. Any RTC resistor connected between the TC and RREF pins has no effect on the output voltage setting at 25C because the TC pin voltage is equal to the RREF regulation voltage at 25C. The RREF resistor value should be approximately 10k because the LT8304 is trimmed and specified using this value. If the RREF resistor value varies considerably from 10k, additional errors will result. However, a variation in RREF up to 10% is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios. First, build and power up the application with the starting RREF, RFB values (no RTC resistor yet) and other components connected, and measure the regulated output voltage, VOUT(MEAS). The new RFB value can be adjusted to: RFB(NEW) = VOUT VOUT(MEAS) *RFB Second, with a new RFB resistor value selected, the output diode temperature coefficient in the application can be tested to determine the RTC value. Still without the RTC resistor, the VOUT should be measured over temperature at a desired target output load. It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT8304. If freeze spray or a heat gun is used, there can be a significant mismatch in temperature between the two devices that causes significant error. Attempting to extrapolate the data from a diode data sheet is another option if there is no method to apply uniform heating or cooling such as an oven. With at least two data points spreading across the operating temperature range, the output diode temperature coefficient can be determined by: - ( VF /T ) = VOUT ( T1) - VOUT ( T2) T1- T2 Using the measured output diode temperature coefficient, an exact RTC value can be selected with the following equation: R TC = (VTC /T ) * RFB - ( VF /T ) NPS Once the RREF, RFB, and RTC values are selected, the regulation accuracy from board to board for a given application will be very consistent, typically under 5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within 1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. 8304fa For more information www.linear.com/LT8304 11 LT8304/LT8304-1 APPLICATIONS INFORMATION Output Power A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter. A boost converter has a relatively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinuous input and output currents which make it similar to a nonisolated buck-boost converter. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 110V during the switch-off time. 40V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 110V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 36V and a maximum input voltage of 75V. A six-to-one winding ratio fits this design example perfectly and outputs equal to 19.0W at 75V but lowers to 14.4W at 36V. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 25 20 N = 12:1 N = 8:1 15 N = 6:1 10 N = 4:1 ASSUME 80% EFFICIENCY 0 20 40 60 INPUT VOLTAGE (V) 80 MAXIMUM OUTPUT POWER 20 OUTPUT POWER (W) N = 2:1 10 N = 1:1 ASSUME 90% EFFICIENCY 0 20 8304 F01 25 MAXIMUM OUTPUT POWER 20 N = 8:1 N = 6:1 15 N =4:1 10 N = 2:1 5 40 60 INPUT VOLTAGE (V) 80 100 8304 F03 Figure 3. Output Power for 12V Output OUTPUT POWER (W) 25 N = 2:1 N = 3:2 N = 1:1 15 10 N = 1:2 5 ASSUME 85% EFFICIENCY 0 20 40 60 INPUT VOLTAGE (V) 80 100 0 ASSUME 90% EFFICIENCY 0 8304 F02 Figure 2. Output Power for 5V Output 12 N = 3:1 15 0 100 Figure 1. Output Power for 3.3V Output 0 N = 4:1 5 5 0 MAXIMUM OUTPUT POWER 20 OUTPUT POWER (W) OUTPUT POWER (W) 25 MAXIMUM OUTPUT POWER 20 40 60 INPUT VOLTAGE (V) 80 100 8304 F04 Figure 4. Output Power for 24V Output 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 APPLICATIONS INFORMATION The equations below calculate output power: the power switch shorter than approximately 160ns. This minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor current exceeds the desired current limit during that time, oscillation may occur at the output as the current control loop will lose its ability to regulate. Therefore, the following equation relating to maximum input voltage must also be followed in selecting primary-side magnetizing inductance: POUT = * VIN * D * ISW(MAX) * 0.5 = Efficiency = ~85% D =Duty Cycle = ( VOUT + VF ) *NPS ( VOUT + VF ) *NPS + VIN ISW(MAX) = Maximum switch current limit = 2A (MIN) LPRI Primary Inductance Requirement The LT8304 obtains output voltage information from the reflected output voltage on the SW pin. The conduction of secondary current reflects the output voltage on the primary SW pin. The sample-and-hold error amplifier needs a minimum 350ns to settle and sample the reflected output voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of 350ns. The following equation gives the minimum value for primary-side magnetizing inductance: LPRI tON(MIN) * VIN(MAX) ISW(MIN) tON(MIN) = Minimum switch-on time = 160ns (TYP) In general, choose a transformer with its primary magnetizing inductance about 40% to 60% larger than the minimum values calculated above. A transformer with much larger inductance will have a bigger physical size and may cause instability at light load. Selecting a Transformer tOFF(MIN) *NPS * ( VOUT + VF ) Transformer specification and design is perhaps the most critical part of successfully applying the LT8304. In addition to the usual list of guidelines dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. ISW(MIN) tOFF(MIN) = Minimum switch-off time = 350ns (TYP) ISW(MIN) = Minimum switch current limit = 0.48A (TYP) Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT8304. Table 1 shows the details of these transformers. In addition to the primary inductance requirement for the minimum switch-off time, the LT8304 has minimum switch-on time that prevents the chip from turning on Table 1. Predesigned Transformers - Typical Specifications TRANSFORMER PART NUMBER DIMENSION (W x L x H) (mm) LPRI (H) TYP LLKG (H) TYP (MAX) TARGET APPLICATION NP:NS VENDOR VIN (V) VOUT (V) IOUT (A) 750315125 17.75 x 13.46 x 12.70 40 1 (2) 6:1 Wurth Elektronik 36 - 75 5 3 750315126 17.75 x 13.46 x 12.70 40 0.5 (1) 2:1 Wurth Elektronik 36 - 75 12 1.2 750315835 17.75 x 13.46 x 12.70 40 1 (2) 8:1 Wurth Elektronik 36 - 75 3.3 4.2 750315836 17.75 x 13.46 x 12.70 40 0.45 (0.9) 1:1 Wurth Elektronik 36 - 75 24 0.6 750315837 17.75 x 13.46 x 12.70 40 0.5 (1) 1:2 Wurth Elektronik 36 - 75 48 0.3 750315839 17.75 x 13.46 x 12.71 40 0.25 (0.5) 1:10 Wurth Elektronik 4 - 36 200 0.012 13324-T083 18.0 x 13.5 x 12.5 40 (2) 8:1 Sumida 36 - 75 3.3 4.2 13324-T084 18.0 x 13.5 x 12.5 40 (1.2) 1:1 Sumida 36 - 75 24 0.6 13324-T085 18.0 x 13.5 x 12.5 40 (1.2) 1:2 Sumida 36 - 75 48 0.3 13324-T086 18.0 x 13.5 x 12.6 40 (1.2) 1:5 Sumida 4 - 36 200 0.012 13324-T087 18.0 x 13.5 x 12.5 40 (1.2) 1:10 Sumida 4 - 18 400 0.006 8304fa For more information www.linear.com/LT8304 13 LT8304/LT8304-1 APPLICATIONS INFORMATION Turns Ratio Note that when choosing an RFB/RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, the use of simple ratios of small integers, e.g., 3:1, 2:1, 1:1, etc., provides more freedom in settling total turns and mutual inductance. Typically, choose the transformer turns ratio to maximize available output power. For low output voltages (3.3V or 5V), a N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer's current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 150V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure NPS < 150V - VIN(MAX) - VLEAKAGE VOUT + VF For larger N:1 step-down turns ratio, choose a transformer with a larger physical size to deliver additional current. In addition, choose a large enough inductance value to ensure that the switch-off time is long enough to accurately sample the output voltage. Always choose the LT8304 for N:1 step-down transformer turns ratio. For lower output power levels or higher output voltage, choose a 1:1 or 1:N step-up transformer for the absolute smallest transformer size. A 1:N step-up transformer will minimize the magnetizing inductance and size, but will also limit the available output power. A higher 1:N step-up turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. For any 1:N step-up transformer turns ratio larger than or equal to 5, the LT8304-1 is recommended. 14 The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within 1%. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8304, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8304. Leakage Inductance and Snubbers Transformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 110V. This leaves at least 40V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 APPLICATIONS INFORMATION <150V VLEAKAGE <110V VSW tOFF > 350ns tSP < 250ns TIME 8304 F05 Figure 5. Maximum Voltages for SW Pin Flyback Waveform In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trigger boundary mode detector, the LT8304 internally blanks the boundary mode detector for approximately 250ns. Any remaining voltage ringing after 250ns may turn the power switch back on again before the secondary current falls to zero. In this case, the LT8304 enters continuous conduction mode. So the leakage inductance spike ringing should be limited to less than 250ns. To clamp and damp the leakage voltage spikes, a (RC + DZ) snubber circuit in Figure 6 is recommended. The RC (resistor-capacitor) snubber quickly damps the voltage spike ringing and provides great load regulation and EMI performance. And the DZ (diode-Zener) ensures well defined and consistent clamping voltage to protect SW pin from exceeding its 150V absolute maximum rating. L Z D * C R * then add capacitance until the period of the ringing is 1.5 to 2 times longer. The change in period determines the value of the parasitic capacitance, from which the parasitic inductance can be also determined from the initial period. Once the value of the SW node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically damp the ringing. The equation for deriving the optimal series resistance using the observed periods ( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is: CPAR = LPAR = CSNUBBER tPERIOD(SNUBBED) 2 -1 tPERIOD 2 tPERIOD CPAR * 4 2 RSNUBBER = LPAR CPAR Note that energy absorbed by the RC snubber will be converted to heat and will not be delivered to the load. In high voltage or high current applications, the snubber needs to be sized for thermal dissipation. A 220pF capacitor in series with a 100 resistor is a good starting point. For the DZ snubber, proper care should be taken when choosing both the diode and the Zener diode. Schottky diodes are typically the best choice, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch voltage protection. The best compromise is to choose the largest voltage breakdown with 5V margin. Use the following equation to make the proper choice: VZENNER(MAX) 145V - VIN(MAX) 8304 F06 Figure 6. (RC + DZ) Snubber Circuit The recommended approach for designing an RC snubber is to measure the period of the ringing on the SW pin when the power switch turns off without the snubber and For an application with a maximum input voltage of 80V, choose a 62V Zener diode, the VZENER(MAX) of which is around 65V. The power loss in the DZ snubber determines the power rating of the Zener diode. A 1.5W Zener diode is typically recommended. 8304fa For more information www.linear.com/LT8304 15 LT8304/LT8304-1 APPLICATIONS INFORMATION Undervoltage Lockout (UVLO) A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO enable falling threshold is set at 1.214V with 14mV hysteresis. In addition, the EN/UVLO pin sinks 2.5A when the voltage on the pin is below 1.214V. This current provides user programmable hysteresis based on the value of R1. The programmable UVLO thresholds are: 1.228V * (R1+R2) +2.5A *R1 R2 1.214V * (R1+R2) VIN(UVLO- ) = R2 VIN(UVLO+ ) = Figure 7 shows the implementation of external shutdown control while still using the UVLO function. The NMOS grounds the EN/UVLO pin when turned on, and puts the LT8304 in shutdown with quiescent current less than 3A. R1 EN/UVLO RUN/STOP CONTROL (OPTIONAL) R2 GND 8304 F07 Figure 7. Undervoltage Lockout (UVLO) Minimum Load Requirement The LT8304 samples the isolated output voltage from the primary-side flyback pulse waveform. The flyback pulse occurs once the primary switch turns off and the secondary winding conducts current. In order to sample the output voltage, the LT8304 has to turn on and off for a minimum amount of time and with a minimum frequency. The LT8304 delivers a minimum amount of energy even 16 ILOAD(MIN) = LPRI *ISW(MIN)2 * fMIN 2 * VOUT LPRI = Transformer primary inductance ISW(MIN) = Minimum switch current limit = 0.53A (MAX) fMIN = Minimum switching frequency = 14kHz (MAX) The LT8304 typically needs less than 0.5% of its full output power as minimum load. Alternatively, a Zener diode with its breakdown of 10% higher than the output voltage can serve as a minimum load if pre-loading is not acceptable. For a 5V output, use a 5.6V Zener with cathode connected to the output. The LT8304-1 requires slightly higher minimum load, typically 2% of full load. Output Short Protection VIN LT8304 during light load conditions to ensure accurate output voltage information. The minimum energy delivery creates a minimum load requirement, which can be approximately estimated as: When the output is heavily overloaded or shorted to ground, the reflected SW pin waveform rings longer than the internal blanking time. After the 350ns minimum switch-off time, the excessive ringing falsely triggers the boundary mode detector and turns the power switch back on again before the secondary current falls to zero. Under this condition, the LT8304 runs into continuous conduction mode at 350kHz (TYP) maximum switching frequency. If the sampled RREF voltage is still less than 0.6V after 11ms (typ) soft-start timer, the LT8304 initiates a new soft-start cycle. If the sampled RREF voltage is larger than 0.6V after 11ms, the switch current may run away and exceed the 2.4A maximum current limit. Once the switch current hits 3.6A over current limit, the LT8304 also initiates a new soft-start cycle. Under either condition, the new soft-start cycle throttles back both the switch current limit and switch frequency. The output short-circuit protection prevents the switch current from running away and limits the average output diode current. 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 APPLICATIONS INFORMATION Design Example Step 2: Determine the primary inductance. Use the following design example as a guide to designing applications for the LT8304. The design example involves designing a 5V output with a 2.8A load current and an input range from 36V to 75V. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 75V, VOUT = 5V, IOUT = 2.8A Step 1: Select the transformer turns ratio. NPS < LPRI LPRI tOFF(MIN) *NPS * ( VOUT + VF ) ISW(MIN) tON(MIN) * VIN(MAX) 150V - VIN(MAX) - VLEAKAGE VOUT + VF VLEAKAGE = Margin for transformer leakage spike = 40V VF = Output diode forward voltage = ~0.3V Example: 150V - 75V - 40V NPS < = 6.6 5V +0.3V ISW(MIN) tOFF(MIN) = 350ns tON(MIN) = 160ns ISW(MIN) = 0.48A Example: 350ns * 6 * ( 5V +0.3V ) = 23H 0.48A 160ns * 75V LPRI = 25H 0.48A LPRI The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 2 shows the switch voltage stress and output current capability at different transformer turns ratio. Table 2. Switch Voltage Stress and Output Current Capability vs Turns Ratio NPS VSW(MAX) at VIN(MAX) (V) IOUT(MAX) at VIN(MIN) (A) DUTY CYCLE (%) 4:1 96.2 2.27 22 to 37 5:1 101.5 2.59 26 to 42 6:1 106.8 2.87 30 to 47 Clearly, only NPS = 6 can meet the 2.8A output current requirement, so NPS = 6 is chosen as the turns ratio in this example. Most transformers specify primary inductance with a tolerance of 20%. With other component tolerance considered, choose a transformer with its primary inductance 40% to 60% larger than the minimum values calculated above. LPRI = 40H is then chosen in this example. The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 2.8A is necessary to work with the LT8304. The 750315125 from Wurth is chosen as the flyback transformer. 8304fa For more information www.linear.com/LT8304 17 LT8304/LT8304-1 APPLICATIONS INFORMATION Step 3: Choose the output diode. Two main criteria for choosing the output diode include forward current rating and reverse-voltage rating. The maximum load requirement is a good first-order guess at the average current requirement for the output diode. Under output short-circuit condition, the output diode needs to conduct much higher current. Therefore, a conservative metric is 60% of the maximum switch current limit multiplied by the turns ratio: IDIODE(MAX) = 0.6 * ISW(MAX) * NPS IDIODE(MAX) = 8.6A The snubber circuit protects the power switch from leakage inductance voltage spike. A (RC + DZ) snubber is recommended for this application. A 220pF capacitor in series with a 100 resistor is chosen as the RC snubber. VZENNER(MAX) 145V - VIN(MAX) Next calculate reverse voltage requirement using maximum VIN: VIN(MAX) The PDS835L (8A, 35V diode) from Diodes Inc. is chosen. Step 4: Choose the output capacitor. VREVERSE > VSW(MAX) VSW(MAX) = VIN(MAX) + VZENNER(MAX) Example: The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the following equation to calculate the output capacitance: LPRI *ISW 2 2 * VOUT * VOUT VREVERSE > 150V A 150V, 1A diode from Diodes Inc. (DFLS1150) is chosen. Step 6: Select the RREF and RFB resistors. Use the following equation to calculate the starting values for RREF and RFB: Example: Design for output voltage ripple less than 1% of VOUT, i.e., 100mV. 2 VZENNER(MAX) 145V - 75V = 70V Choose a diode that is fast and has sufficient reverse voltage breakdown: 75V VREVERSE = 5V + = 17.5V 6 40H * ( 2.4A ) COUT = = 230F 2 * 5V * 0.1V Example: A 62V Zener with a maximum of 65V will provide optimal protection and minimize power loss. So a 62V, 1.5W Zener from Central Semiconductor (CMZ5944B) is chosen. NPS Example: COUT = Step 5: Design snubber circuit. The maximum Zener breakdown voltage is set according to the maximum VIN: Example: VREVERSE = VOUT + Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop up to 40% of quoted capacitance at the maximum voltage rating. So three 100F, 10V rating ceramic capacitors are chosen. RFB = ( ) RREF * NPS * VOUT + VF ( TO) VREF RREF = 10k Example: RFB = 10k * 6 * ( 5V +0.3V ) = 318k 1.00V For 1% standard values, a 316k resistor is chosen. 18 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 APPLICATIONS INFORMATION Step 7: Adjust RFB resistor based on output voltage. Step 9: Select the EN/UVLO resistors. Build and power up the application with application components and measure the regulated output voltage. Adjust RFB resistor based on the measured output voltage: Determine the amount of hysteresis required and calculate R1 resistor value: RFB(NEW) = VOUT VOUT(MEASURED) Example: *RFB Choose 2.5V of hysteresis, R1 = 1M Example: RFB = VIN(HYS) = 2.5A * R1 Determine the UVLO thresholds and calculate R2 resistor value: 5V * 316k = 309k 5.11V Step 8: Select RTC resistor based on output voltage temperature variation. VIN(UVLO+) = 1.228V * (R1+ R2) + 2.5A * R1 R2 Example: Measure output voltage in a controlled temperature environment like an oven to determine the output temperature coefficient. Measure output voltage at a consistent load current and input voltage, across the operating temperature range. Set VIN UVLO rising threshold to 34.5V: Calculate the temperature coefficient of VF: Step 10: Ensure minimum load. VOUT ( T1) - VOUT ( T2) T1- T2 3.35mV/C RFB R TC = * - ( VF /T ) NPS The theoretical minimum load can be approximately estimated as: - ( VF /T ) = 2 40H * ( 0.53A ) *14kHz ILOAD(MIN) = = 15.7mA 2 * 5V Example: - ( VF /T ) = R TC = 5.149V - 4.977V = 1.72mV / C 100C - ( 0C) 3.35mV/C 309 * = 100k 1.72mV/C 6 R2 = 40.2k VIN(UVLO+) = 34.3V VIN(UNLO-) = 31.4V Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the converter delivers more energy than what is consumed at the output. The real minimum load for this application is about 20mA. In this example, a 249 resistor is selected as the minimum load. 8304fa For more information www.linear.com/LT8304 19 LT8304/LT8304-1 TYPICAL APPLICATIONS 18V to 80VIN/3.3VOUT Isolated Flyback Converter T1 8:1 C3 220pF * 0.63H R3 40H 100 * Z1 R1 1M D1 VIN R2 88.7k C2 1F SW EN/UVLO R4 274k LT8304 RFB GND INTVCC C4 330F x2 RREF R6 100k TC VOUT- D1: DIODES DFLS1150 D2: DIODES SBR15U30SP5 T1: SUMIDA 13324-T083 Z1: CENTRAL CMZ5944B R5 10k 100 VOUT+ 3.3V 25mA TO 3.4A (VIN = 24V) 25mA TO 4.8A (VIN = 48V) 25mA TO 5.6A (VIN = 72V) 90 EFFICIENCY (%) D2 VIN 18V TO 80V C1 10F Efficiency vs Load Current 80 70 60 VIN = 24V VIN = 48V VIN = 72V 50 8304 TA02a 40 0 0.8 1.6 2.4 3.2 4.0 LOAD CURRENT (A) 4.8 5.6 8304 TA02b 18V to 80VIN/5VOUT Isolated Flyback Converter VIN 18V TO 80V Z1 C1 10F R1 1M R2 88.7k C2 1F D1 VIN EN/UVLO SW R4 309k LT8304 GND INTVCC RFB RREF TC T1 6:1 C3 220pF * 1.1H R3 40H 100 * R6 100k R5 10k VOUT+ 5V 20mA TO 2.4A (VIN = 24V) 20mA TO 3.6A (VIN = 48V) C4 20mA TO 4.2A (VIN = 72V) 100F x3 VOUT- D1: DIODES DFLS1150 D2: DIODES PDS835L T1: WURTH 750315125 Z1: CENTRAL CMZ5944B 100 90 EFFICIENCY (%) D2 Efficiency vs Load Current 80 70 60 VIN = 24V VIN = 48V VIN = 72V 50 8304 TA03a 40 0 0.6 1.2 1.8 2.4 3.0 LOAD CURRENT (A) 3.6 4.2 8304 TA03b 20 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 TYPICAL APPLICATIONS 18V to 80VIN/12VOUT Isolated Flyback Converter Z1 C1 10F R1 1M R2 88.7k C2 1F D1 VIN EN/UVLO SW RFB INTVCC RREF TC * 10H C4 47F R6 OPEN R5 10k 100 VOUT+ 12V 10mA TO 1.0A (VIN = 24V) 10mA TO 1.4A (VIN = 48V) 10mA TO 1.6A (VIN = 72V) VOUT- R4 237k LT8304 GND C3 220pF R3 40H 100 * D2 D1: DIODES DFLS1150 D2: DIODES PMEG6030EP T1: WURTH 750315126 Z1: CENTRAL CMZ5944B 90 EFFICIENCY (%) VIN 18V TO 80V T1 2:1 Efficiency vs Load Current 80 70 60 VIN = 24V VIN = 48V VIN = 72V 50 8304 TA04a 40 0 0.4 0.8 1.2 LOAD CURRENT (A) 1.6 8304 TA04b 18V to 80VIN/24VOUT Isolated Flyback Converter Z1 C1 10F R1 1M R2 88.7k C2 1F D1 VIN EN/UVLO SW R4 237k LT8304 GND INTVCC RFB RREF TC C3 220pF R3 40H 100 * R6 OPEN R5 10k T1 1:1 D2 * 40H C4 10F 100 VOUT+ 24V 5mA TO 0.5A (VIN = 24V) 5mA TO 0.7A (VIN = 48V) 5mA TO 0.8A (VIN = 72V) VOUT- D1: DIODES DFLS1150 D2: DIODES SBR2U150SA T1: SUMIDA 13324-T084 Z1: CENTRAL CMZ5944B 90 EFFICIENCY (%) VIN 18V TO 80V Efficiency vs Load Current 80 70 60 VIN = 24V VIN = 48V VIN = 72V 50 8304 TA05a 40 0 0.1 0.2 0.3 0.4 0.5 0.6 LOAD CURRENT (A) 0.7 0.8 8304 TA05b 8304fa For more information www.linear.com/LT8304 21 LT8304/LT8304-1 TYPICAL APPLICATIONS 18V to 80VIN/48VOUT Isolated Flyback Converter Efficiency vs Load Current VIN 18V TO 80V Z1 C1 10F R1 1M R2 88.7k C2 1F D1 VIN EN/UVLO SW INTVCC RFB RREF TC C4 2.2F R4 232k LT8304 GND T1 1:2 C3 220pF * 160H R3 40H 100 * R6 OPEN R5 10k 100 VOUT+ 48V 2mA TO 0.24A (VIN = 24V) 2mA TO 0.34A (VIN = 48V) 2mA TO 0.40A (VIN = 72V) VOUT- D1: DIODES DFLS1150 D2: DIODES SBR1U400P1 T1: SUMIDA 13324-T085 Z1: CENTRAL CMZ5944B 90 EFFICIENCY (%) D2 80 70 60 VIN = 24V VIN = 48V VIN = 72V 50 8304 TA06a 40 0 0.1 0.2 0.3 LOAD CURRENT (A) 0.4 8304 TA06b 22 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 TYPICAL APPLICATIONS 4V to 36VIN/200VOUT Isolated Flyback Converter T1 1:5 VIN 4V TO 36V 10F 50V 40H VIN EN/UVLO SW 1F 6.3V * 1mH * VOUT- RFB INTVCC 0.33F 250V 392k LT8304-1 TC VOUT+ 200V 0.3mA TO 12mA (VIN = 4V) 0.5mA TO 35mA (VIN = 12V) 1.5mA TO 75mA (VIN = 36V) D1 T1: SUMIDA 13324-T086 D1: CENTRAL CMMR1U-06 TR RREF 10k GND 10pF 8304 TA07a Efficiency, VOUT = 200V 95 Load Regulation, VOUT = 200V Efficiency, V OUT = 200V 210 Load Regulation, VOUT = 200V 90 OUTPUT VOLTAGE (V) EFFICIENCY (%) 85 80 75 70 VIN = 4V VIN = 12V VIN = 36V 65 60 0 15 30 45 60 LOAD CURRENT (mA) 75 205 200 195 190 8304 TA07b VIN = 4V VIN = 12V VIN = 36V 0 15 30 45 60 LOAD CURRENT (mA) 75 8304 TA07c 8304fa For more information www.linear.com/LT8304 23 LT8304/LT8304-1 4V to 18VIN/400VOUT Isolated Flyback Converter T1 1:10 VIN 4V TO 18V 10F 25V * 40H VIN EN/UVLO SW 1F 6.3V 4mH * 0.15F 600V VOUT- 392k LT8304-1 TC VOUT+ 400V 0.4mA TO 6mA (VIN = 4V) 0.4mA TO 20mA (VIN = 12V) 0.4mA TO 30mA (VIN = 18V) D1 RFB T1: SUMIDA 13324-T087 D1: CENTRAL CMMR1U-06 TR RREF INTVCC 10k GND 10pF 8304 TA08a Efficiency, VOUT = 400V Load Regulation, VOUT = 400V OUT 90 420 OUTPUT VOLTAGE (V) EFFICIENCY (%) 80 70 60 50 40 VIN = 4V VIN = 12V VIN = 18V 0 5 10 15 20 LOAD CURRENT (mA) 25 30 410 400 390 380 8304 TA08b 24 Load Regulation, VOUT = 400V VIN = 4V VIN = 12V VIN = 18V 0 5 10 15 20 LOAD CURRENT (mA) 25 30 8304 TA08c 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 TYPICAL APPLICATIONS -4V to -80VIN/12VOUT Buck-Boost Converter VIN R4 Z1 121k SW RFB EN/UVLO C1 10F VOUT +12V 5mA TO 0.25A (VIN = -5V) 5mA TO 0.7A (VIN = -24V) 5mA TO 0.8A (VIN = -48V) C3 47F 5mA TO 0.9A (VIN = -72V) D1 LT8304 RREF INTVCC C2 1F D1: DIODES SBR2U150SA L1: WURTH 744771133 Z1: CENTRAL CMHZ5243B R5 10k GND VIN -4V TO -80V 100 90 EFFICIENCY (%) L1 33H Efficiency vs Load Current 80 70 60 VIN = -5V VIN = -24V VIN = -48V VIN = -72V 50 8304 TA09a 40 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 LOAD CURRENT (A) 8304 TA09b -18V to -80VIN/-12VOUT Negative Buck Converter Efficiency vs Load Current 100 C3 47F R1 806k C1 10F R2 71.5k VIN EN/UVLO INTVCC VIN -18V TO -80V SW LT8304 EN/UVLO C2 1F L1 33H 90 VOUT -12V 10mA TO 1A R4 121k D1: DIODES SBR2U150SA L1: WURTH 744771133 Z1: CENTRAL CMHZ5243B RFB RREF R5 10k EFFICIENCY (%) Z1 D1 80 70 60 40 8304 TA10a VIN = -24V VIN = -48V VIN = -72V 50 0 0.2 0.4 0.6 LOAD CURRENT (A) 0.8 1.0 8304 TA10b 8304fa For more information www.linear.com/LT8304 25 LT8304/LT8304-1 PACKAGE DESCRIPTION Please refer to http://www.linear.com/product/LT8304#packaging for the most recent package drawings. S8E Package 8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad (Reference LTC DWG # 05-08-1857 Rev C) .050 (1.27) BSC .189 - .197 (4.801 - 5.004) NOTE 3 .045 .005 (1.143 0.127) 8 .089 .160 .005 (2.26) (4.06 0.127) REF .245 (6.22) MIN .150 - .157 .080 - .099 (2.032 - 2.530) (3.810 - 3.988) NOTE 3 .228 - .244 (5.791 - 6.197) 1 .030 .005 (0.76 0.127) TYP .005 (0.13) MAX 7 5 6 .118 (2.99) REF 3 2 .118 - .139 (2.997 - 3.550) 4 RECOMMENDED SOLDER PAD LAYOUT .010 - .020 x 45 (0.254 - 0.508) .008 - .010 (0.203 - 0.254) .053 - .069 (1.346 - 1.752) 0- 8 TYP .016 - .050 (0.406 - 1.270) .014 - .019 (0.355 - 0.483) TYP NOTE: 1. DIMENSIONS IN INCHES (MILLIMETERS) 2. DRAWING NOT TO SCALE 3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm) 26 4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542) 5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542) 4 5 .004 - .010 0.0 - 0.005 (0.101 - 0.254) (0.0 - 0.130) .050 (1.270) BSC S8E 1015 REV C 8304fa For more information www.linear.com/LT8304 LT8304/LT8304-1 REVISION HISTORY REV DATE DESCRIPTION A 02/17 Added LT8304-1 and H-Grade options Changed TC Pin Current conditions Changed TC pin description to C Added High Step-Up VOUT Applications section Updated Predesigned Transformers - Typical Specifications table Revised Turns Ratio section Added new application circuits and graphs PAGE NUMBER All 3 7 9 13 14 23, 24 8304fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. For more information www.linear.com/LT8304 27 LT8304/LT8304-1 TYPICAL APPLICATION 4V to 100VIN/140VOUT Boost Converter L1 150H VIN 4V TO 100V VIN SW RFB EN/UVLO C1 10F VOUT 140V 1.5mA TO 25mA (VIN = 5V) 2mA TO 300mA (VIN = 48V) 7mA TO 700mA (VIN = 100V) D1 R3 1M LT8304 RREF INTVCC C2 1F C3 1F R4 Z1 499k Z2 D1: DIODES DFLS1200 L1: COILCRAFT DS5022P-154MLB Z1, Z2: CENTRAL CMHZ5207B R5 3.57k GND 8304 TA11a Efficiency vs Load Current 100 EFFICIENCY (%) 90 80 70 60 VIN = 5V VIN = 48V VIN = 100V 50 40 1 10 100 LOAD CURRENT (mA) 1000 8304 TA11b RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E LT8303 100VIN Micropower Isolated Flyback Converter with 150V/450mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V VCC 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573/LT3574 LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V VIN 100V, No-Opto Flyback, MSOP-16(12) LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3757A/LT3759/ LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch 28 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT8304 (408) 432-1900 FAX: (408) 434-0507 www.linear.com/LT8304 8304fa LT 0217 REV A * PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2016